Preliminary cascade unch. Classification of amplification devices

Pre-amplification stages. The typical signal source used to develop output voltage at the level of 50-200 mV. High-quality amplifiers were oriented towards this voltage. Correction circuits were previously located between the input sockets and the grid of the first lamp, in which the signal was attenuated by at least half (6 dB) at the most sensitive input. In the fine-compensated volume control, the minimum signal attenuation is another 6 dB. Tone controls that provide ±20dB of control typically attenuate the signal by another 30-40dB. If there were cathode followers in the input circuits, the signal loss increased by another 3-6 dB. So, the total signal attenuation used to be 45-58 dB. The signal voltage on the grids of the final stage lamps averages 10-20 V. The ratio of this value to the input signal voltage is 10/0.05 = 200 (46 dB). So, the amplification of the preliminary stages, taking into account the signal attenuation and the required voltage on the grids of the final stage lamps, should previously have been on the order of 90-100 dB. In other words, the gain of the preliminary stages should be approximately 100,000. This is quite a significant value for a low-frequency amplifier. If the voltage gain of each of the amplifier stages is approximately 10, then, obviously, the number of stages should be equal to 5. If the gain of each stage is about 100, the total number of stages will be equal to 3 (with some margin). Since a gain of 10 per stage is provided by almost any modern low-frequency tube triode, and a gain of 100 per stage is the limit even for good low-frequency pentodes, it can be argued that for tube amplifiers the number of pre-amplification stages should range from three until five.

How many cascades should you make: 3 or 5? The first answer, of course, is “3”. However, there is no need to rush. Three cascades - this means the minimum gain of the cascade is equal to the third root of 10000. Note that this is not the μ of the lamp, but the gain of the cascade, which rarely exceeds 50% of the μ of the lamp. Therefore, triodes are no longer needed. This means there will be three cascades on pentodes or, in extreme cases, two on pentodes and one on a triode. The latter circuit, which does not have any gain margin, does not allow the use of negative feedback in the circuit, i.e. practically unsuitable for Hi-Fi amplifiers, because without a negative feedback It is unthinkable to reduce the nonlinear distortion factor and expand the frequency range to the required values. Three stages on pentodes can allow the introduction of negative feedback, but then the first, input stage is also assembled on the pentode, and in this case, as experience shows, it is almost impossible to achieve a complete absence of microphone effect and a background level below 60 dB. The other extreme - five stages on triodes - always provides the required gain even on the worst tubes, however, using tubes with an average gain of about 20-50, it is easy to obtain the required gain with a sufficient margin with four triodes (i.e. on two double lamps). This scheme is the most common. True, many foreign companies produce a specially designed pentode for the input stage with a low level of self-noise and not prone to microphone effects (EF-184, EF-804, etc.). Using such a pentode and subsequent triodes with a large μ (90-120) of the ECC-83 type, it is possible to obtain the required gain on three stages using the pentode - triode - triode system, but firstly, such a system requires the use of special lamps, and - secondly - very high quality transformer steel, highly sensitive end lamps, etc. Therefore, this scheme is not suitable.

Note. In the 21st century, the situation has changed significantly. Nowadays no one is using physical analogue pre-amplifier stages. Pre-processing of the signal is trusted to high-quality DACs. The input signal is considered normal at 1-2 volts. Therefore, for a tube terminal, an amplification of 20-50 times is sufficient. And only one can cope with this task electric lamp in the pre-amp stage. This is, for example, a double triode, which combines the functions of a bass reflex. That is why all the garbage from numerous successive cascades remains in the distant past. Evgeny Bortnik.

Bass reflexes. If the phase inverter is assembled according to a circuit in which each arm is also an amplifier (for example, according to the circuit in Fig. 1), then the gain of this arm is taken into account in the overall gain of the path. We remind you that you need to take into account the gain of only one arm, since the second arm of the inverter is only a matcher for the second arm of the push-pull final stage and is not part of the general amplification path.

If the phase inverter is assembled according to a symmetrical cathode follower circuit (Fig. 2), then its gain is always less than unity, so such a stage is not only not an amplification stage, but also requires an additional increase in the total gain by 4-6 dB.

The method for selecting the gain for a transistor amplifier is exactly the same. Now specifically about the circuits of the pre-amplifier stages themselves. These are the simplest resistive amplifiers without any circuit features. Typical for all stages, both triodes and pentodes, are the anode (collector) loads reduced by 2-5 times compared to the optimal calculated values ​​for expanding the bandwidth towards higher frequencies, increased to 0.1-0. 25 µF transition capacitors and up to 1-1.5 MΩ grid leakage resistors to reduce frequency response roll-off by low frequencies ah, the use of negative current feedback in all stages, except for the one on which the frequency response control unit is assembled. As for the amplification elements themselves, in recent years many different new types of lamps and transistors with excellent parameters have appeared. Thus, the value of S for low-power lamps became equal to 30-50 mA/V against the usual values ​​of 3-10 mA/V, and therefore the sensitivity of the lamps increased sharply. Calculations show that theoretically all the pre-amplification can be obtained even on two stages with such lamps. However, it would be useful to warn amateurs against haste in choosing such lamps. And the point here is not conservatism, but the fact that an increase in, say, the slope of the lamps is achieved by a sharp decrease in the gap between the control grid and the cathode, which significantly increases the tendency of the lamp to generate thermal currents and the resulting huge nonlinear distortions. Also important are the high cost and lower durability of such lamps. It can be argued that such tubes as 6N1P, 6N2P, 6NZP, 6N23P, 6N24P, 6Zh1P, 6Zh5P, proven by many years of practice, are quite suitable for the preliminary stages of even the best, most modern amplifiers. For example, below are shown several circuits of the CPU on lamps in their normal modes

In Fig.3. tube pre-amp stages are shown. a - two-stage amplifier with interstage internal feedback; b - cascade with linearizing feedback in the protective grid circuit.

Final and pre-final stages – power amplifiers. Formally, pre-terminal cascades (drivers, from the English word drive - excite, set, swing) are classified as voltage amplifiers, i.e., preliminary cascades, but they are discussed in this, and not in the previous paragraph, in order to emphasize that by the nature of the work and In terms of modes of use, drivers are much closer to final amplifiers, i.e. power amplifiers. Hi-Fi amplifiers are characterized by a significant output power of the order of 15-50W. This means that to excite (drive) the final stage without noticeable nonlinear distortions, a power of the order of 1-5 W is already required, at a voltage of up to 25-35 V, and if we take into account the requirements for reducing nonlinear distortions, it becomes clear that conventional low-power triodes cannot provide excitation of powerful terminal lamps. Therefore, it becomes logical and justified to use high-power lamps in the last voltage amplification stage. It is possible that, theoretically, it would be more correct to make the pre-final stages in all cases transformer or choke in order to obtain the greatest value utilization factor based on anode voltage ξ, but there are several reasons why this should not be done. The transformer cascade always introduces noticeable frequency distortions, and at powers above 1-2 W, noticeable nonlinear distortions. In addition, transformers are relatively expensive, complex and labor-intensive to manufacture, heavy and bulky, sensitive to magnetic interference and at the same time a source of audio frequency interference for other amplifier circuits (primarily input ones).

At the same time, radio amateurs now have medium-power, broadband and economical lamps at their disposal, which make it possible to easily obtain undistorted power of about 2-4 W at an active load resistance. These primarily include lamps of types 6P15P, 6E5P, 6F3P, 6F4P, 6F5P, 6Zh5P, 6Zh9P, etc. However, this issue needs to be approached more carefully. In some cases, for reasons of simpler coordination, it is still advisable to use a transformer connection. Pre-amplifier circuits are shown below

For final low-frequency cascades with a power of up to 10-12 W, radio amateurs in most cases use 6P14P type lamps, partly because they quite easily provide the specified power. In addition, unfortunately, there are no other lamps suitable for this purpose. Such an outdated, although very good lamp, like 6P3S (6L6) Nowadays It cannot be recommended, and the industry does not produce more powerful special lamps for ULF final stages like the German EL-34. [Strange conclusion, without any reason, in 1980-90 the use of 6P3S cannot be recommended! Pure voluntarism from the Soviet of Deputies. In the 21st century, for example, 6P3S lamps can be strongly recommended for designing a tube amplifier. It is important to find specimens in good preservation. E.B.] People often try to get more power from the same 6P14P tubes by forcing the mode, but this path is completely unacceptable due to the sharp deterioration in the reliability of the amplifier and the increase in nonlinear distortions when a grid thermal current appears.

Taking into account the above, we can recommend that radio amateurs use 6P14P lamps in any push-pull circuits only at powers not exceeding 10 watts. [An amazingly meaningless recommendation in the style of “since there is nothing good, then do what you do.” The author seems to be a cool authority, but he writes nonsense. E.B.] With a higher output power, it is necessary to switch to such obviously not “low-frequency” lamps as 6P31S, 6P36S, 6P20S, GU-50, 6N13S (6N5S) both in classic push-pull and ultra-linear circuits, and in bridge circuits less familiar to radio amateurs circuits, also called push-pull-parallel. The first three of these lamps are intended for use in the final cascades of horizontal scanning televisions and allow you to extract power up to 25 W from two lamps; a GU-50 generator lamp with an anode voltage of 500-750 V (and according to its passport it has Ua.work = 1000 V) is easy gives to push-pull circuit power 40-60W; double triode 6N13S, designed specifically as a control lamp in electronic voltage stabilizer circuits, has a very low internal resistance and, with a relatively low anode voltage, makes it possible to obtain a power of at least 15 W (per one cylinder) in a conventional push-pull circuit, and when switched on, two in each arm triodes in parallel (two cylinders) in conventional push-pull and bridge circuits provides output power up to 25W Using the listed lamps, the radio amateur has a large selection for creative activity.

[Another recommendation in a vague state of consciousness. I wonder why twin or triple lamps are not suitable for creative activities? Maybe the author simply does not know the rules for parallel connection of radioelements? Namely, a parallel connection, with a high-quality selection of copies, gives a lot of intermediate options, very powerful amplifiers with decent characteristics. It is strange to read the recommendation of a 6P31S lamp, which is not at all more powerful than 6P14P, but is much worse in characteristics. And it’s also disappointing to see quick recommendations for the use of 6N13C lamps (paralleled, by the way). An amazing demonstration of frivolity, since the author is completely unaware of practice, because 6N13C lamps are rare guano. The spread of characteristics of the halves has a range of 100% or more. It is almost impossible to accurately select them for parallel connection, so the amplifier cannot deliver significant power to the load without overheating one of the halves, and the utilization factor is unlikely to exceed 40-50%. AND simple circuits parallel connection for 6N13S, without leveling body kits, are unsuitable. And the discussions about lamps are touching, because there are a large number of other excellent lamps, in contrast to the recommended ones, for example 6P13S, 6P44S, 6P45S, G807; in extreme cases, 6P3S lamps are suitable. E.B.]

Fig.5. Powerful final stages of the low-frequency ULF path. a - on 6P36S lamps in ultralinear switching; b - on GU-50 lamps in a push-pull parallel circuit; c - on 6N13S lamps with fixed bias balancing

Since all circuits were considered as low-frequency, i.e. designed for a limited bandwidth (no more than 5-8 kHz), nothing was said about output transformers, chokes, and autotransformers. All of them are the most common, assembled on W-shaped or strip cores made of simple transformer steel 0.35 mm thick. There are no increased requirements for the frame design and windings, with the exception of high degree symmetry of the individual halves of the primary winding. This requirement is especially important for ultra-linear circuits for switching terminal lamps. The values ​​of leakage inductance and capacitance of the primary winding are not significant. Secondary windings with powers above 10 W should be wound with as thick a wire as possible to reduce active losses. It is advisable to make several taps to select the best operating mode for the final stage. This issue is discussed in more detail in the next paragraph. The high-frequency final stages of two-channel Hi-Fi amplifiers are significantly different from the low-frequency ones, so the recommendations regarding them will be different. First of all, this applies to the types of lamps. [ Amazing reasoning. The author invented his own classification of LF and HF. Even to a complete amateur who has read the section on vacuum tubes, first of all, it is obvious that the invented frequency division has nothing to do with vacuum tubes at all; their range goes into hundreds of megahertz. The 6P14P lamp is purple, which frequency signals should be amplified, be it 0.1 kHz, 1 kHz, 5 kHz, 8 kHz, 16 kHz or 32 kHz. But with regard to the matching transformer, this question is already relevant. But there is no need to worry here either, because... up to 18-20 kHz, ordinary transformers are suitable; you don’t need to wind anything at all. And for frequencies above 20 kHz you should switch to ferrites. It seems that the author has not heard anything about sectioning the windings to improve the frequency response, and recommends a thick wire for the secondary winding. And the concept of ACTIVE LOSSES is absolute bullshit, since there are no passive losses and there are no reactive losses either. E.B.]

Since the power of high-frequency channels, even in top-class amplifiers, is in the range of 10-12 W, the most suitable lamps are 6P14P and 6N13S. The best switching circuits are push-pull ultralinear, bridged on 6P14P in triode switching, and “two-story” on 6N13S. Regarding the last scheme, the most common version of which is shown in Fig. 6, we can say that although it is not new in a theoretical sense, it became widespread in broadcasting equipment only in the 60s of the last century. As often happens, the scheme has become very widespread, and when talking about the advantages of the scheme, they usually remain silent about its disadvantages. Let's try to objectively evaluate both.

[First of all, I propose to sensibly evaluate the most important consequence of the creation of transformerless circuits. The past 50 years have shown that such schemes have not received any distribution, and could not have received them. As the standard of living rises, the value of health increases. Therefore, the main and insurmountable disadvantage of transformerless circuits is the lack of galvanic isolation from the source high voltage, will never allow such schemes to achieve any kind of spread among the human population. And let the dreamers study and analyze the modes of such circuitry until they are blue in the face.]

Fig.6. One of the most common final stage circuits with lamps connected in series DC

Connecting two lamps in series for direct current is equivalent to the fact that for alternating current both of them are connected in parallel relative to the load, due to which their total internal resistance is actually four times less than that of a conventional push-pull cascade. If for such a circuit we take lamps whose internal resistance is lower than usual, and use relatively high-impedance loudspeakers as a load, then it turns out that the output transformer, according to calculations, would in this case have a transformation coefficient close to unity or, in any case, measured in units. It is then possible to connect the load to the lamps directly, without an output transformer. This, of course, is an unconditional advantage of the scheme. However, this dignity comes at a high price. First of all, direct switching on of the load still turns out to be impossible due to the presence at the points of its switching on of half the voltage of the power source (120-150V). Therefore, the loudspeakers have to be switched on through a decoupling capacitor, the capacitance of which is directly related to the load resistance and the lower limit of the passband. Indeed, if the permissible voltage loss of the useful signal on the separating capacitor is 10% of the value of the signal itself, then at Rн=20 Ohm and flow=40 Hz the reactance of the capacitor should not exceed 2 Ohms, from which its capacitance is equal to

It is clear that only an electrolytic capacitor can have such a capacitance, but it must be remembered that its operating voltage must be at least not lower than the full voltage of the power source, i.e. 300-350V. And then it turns out that the cost of such a capacitor is not at all lower than the cost of the output transformer, especially since, unlike a capacitor, a radio amateur can always make a transformer himself, if necessary. Of course, it is possible to make a loudspeaker with a voice coil resistance of not 20, but 200 Ohms, which, under the same conditions, will allow reducing the capacitance of the coupling capacitor to 200 μF, but in this case the cost of the loudspeaker increases sharply. However, this is not the only drawback of this scheme. The second is that when the lamps are connected in series with direct current, only half the voltage of the anode source is applied to each of them, so the circuit can only work well on special lamps whose rated anode voltage does not exceed 100-150V. However, most lamps of this type have an insignificant maximum output power, rarely exceeding a few watts. In addition, studies have shown that when using pentodes, this circuit is fundamentally somewhat asymmetrical, which makes it unsuitable for the final low-frequency stages of Hi-Fi amplifiers. In high-frequency cascades, the first drawback immediately disappears, since with the values ​​​​selected in the previous calculation and the lower limit of the HF channel flow = 2 kHz, the capacitance value of the separating capacitor

Moreover, in this case, a ten percent signal loss will occur only in the worst, practically non-working part of the passband, and at ftop = 20 kHz the signal loss will be only 1%. In addition, the required output power for the final RF stage is significantly less than for the LF stage, which allows the use of a 6N13C double triode in this circuit, which has low internal resistance and works well at low anode voltages. A practical diagram of such a cascade is shown in Fig. 7.

Fig.7. Practical diagram of a “two-story” final stage based on a double triode 6N13S (6N5S)

If the power of the RF channel does not exceed 2-3W, you can assemble the final stage according to the circuit in Fig. 8 using lamps of types 6F3P or 6F5P. The output transformer for this circuit is assembled on a tape core with a tape thickness of no more than 0.2 mm or on an W-shaped permalloy. In order for the ultralinear circuit to give a noticeable result and for nonlinear distortions to actually be on the order of 0.2-0.5%, the tap point of the primary winding must in each case be selected empirically directly from the results of measurements of the r.n.i. in the process of setting up an amplifier. To do this, when winding a transformer, 4-6 taps must be provided for each half of the primary winding.

Fig.8. Push-pull high-frequency final stage using 6F3P or 6F5P lamps (Pout = 2.5 W)

For transistor amplifiers, the “two-story” circuit, on the contrary, turns out to be preferable to all others. This is explained by the low internal resistance of high-power transistors and collector voltage (compared to lamps). Therefore, excellent matching of the cascade with the load is ensured even when using conventional low-impedance loudspeakers, for example, the 4GD-35 type. In addition, the decoupling capacitor turns out to be small in size even with a capacity of 2000-5000 μF, since its operating voltage does not exceed 20-30V. Such schemes are widespread and well known to radio amateurs.

As a general conclusion, I can cite several considerations that in the 21st century will certainly be perceived as rational. The first consideration is whether it is correct for the author to discuss only push-pull amplifiers, since single-ended circuits are intended for beginners. Secondly, the thoroughness of the approach to systematizing the circuitry of cascades also deserves respect. Third, the author’s indisputable qualifications in some cases border on astounding prejudices, and lapses in thinking are apparently a consequence of the author’s high theoretical preparation and insufficient practical experience. Fourth, the past decades have significantly changed the situation, both in basic concepts and in circuit design, especially with regard to the output stages of high-performance amplifiers. And there is no longer any excessive ceremony. Much has become simpler and clearer. Some show-offs died without showing resilience. But they will be replaced by new show-offs, like oxygen-free copper. It seems very important to realize the fact that changes in the technological structure of society should not change the fundamental life values, for example, Slavic civilization. Prepared a publication based on materials from Gendin’s book downloaded online.

Evgeny Bortnik, Krasnoyarsk, Russia, March 2018

RESISTOR RESEARCH

AMPLIFIER CASCADE

BASIC CONVENTIONS AND ABBREVIATIONS

AFC - amplitude-frequency response;

PH - transient response;

MF - mid frequencies;

LF - low frequencies;

HF - high frequencies;

K is the gain of the amplifier;

Uc is the voltage of the signal with frequency w;

Cp - separation capacitor;

R1,R2 - divider resistance;

Rк - collector resistance;

Re - resistance in the emitter circuit;

Ce - capacitor in the emitter circuit;

Rн - load resistance;

CH - load capacity;

S - transconductor slope;

Lk - correction inductance;

Rf, Sf - elements of low frequency correction.

1. PURPOSE OF THE WORK.

The purpose of this work is:

1) study of the operation of a resistor cascade in the region of low, medium and high frequencies.

2) study of schemes for low-frequency and high-frequency correction of the amplifier’s frequency response;

2. HOMEWORK.

2.1. Study the circuit of a resistor amplifier stage, understand the purpose of all elements of the amplifier and their influence on the parameters of the amplifier (subsection 3.1).

2.2. Learn the operating principle and circuit diagrams low-frequency and high-frequency correction of the amplifier's frequency response (subsection 3.2).

2.3. Understand the purpose of all elements on the front panel of the laboratory layout (section 4).

2.4. Find answers to everything Control questions(section 6).

3. RESISTOR CASCADE ON A BIPOLAR TRANSISTOR

Resistor amplification cascades are widely used in various areas radio engineering. An ideal amplifier has a uniform frequency response over the entire frequency band; a real amplifier always has distortion in the frequency response, primarily a decrease in gain at low and high frequencies, as shown in Fig. 3.1.

AC resistor amplifier circuit for bipolar transistor according to the circuit with a common emitter is shown in Fig. 3.2, where Rc is the internal resistance of the signal source Uc; R1 and R2 - divider resistances that set the operating point of transistor VT1; Re is the resistance in the emitter circuit, which is shunted by the capacitor Se; Rк - collector resistance; Rн - load resistance; Cp - decoupling capacitors that provide DC separation of transistor VT1 from the signal circuit and the load circuit.

The temperature stability of the operating point increases with increasing Re (due to an increase in the depth of negative feedback in the DC cascade), the stability of the operating point also increases with decreasing R1, R2 (due to an increase in the divider current and an increase in the temperature stabilization of the base potential VT1). A possible decrease in R1, R2 is limited by the permissible decrease in the input resistance of the amplifier, and a possible increase in Re is limited by the maximum permissible drop in DC voltage across the emitter resistance.

3.1. Analysis of the operation of a resistor amplifier in the low, medium and high frequencies.

The equivalent circuit was obtained taking into account the fact that on alternating current the power bus (“-E p”) and the common point (“ground”) are short-circuited, and also taking into account the assumption of 1/wCe<< Rэ, когда можно считать эмиттер VT1 подключенным на переменном токе к общей точке.

The behavior of the amplifier is different in the region of low, medium and high frequencies (see Fig. 3.1). At medium frequencies (MF), where the resistance of the coupling capacitor Cp is negligible (1/wCp<< Rн), а влиянием емкости Со можно пренебречь, так как 1/wCо >> Rк, the equivalent circuit of the amplifier is converted into the circuit in Fig. 3.4.

From the diagram in Fig. 3.4 it follows that at medium frequencies the gain of the cascade Ko does not depend on the frequency w:

Ko = - S/(Yi + Yk + Yn),

whence, taking into account 1/Yi > Rн > Rк we obtain the approximate formula

Consequently, in amplifiers with a high-resistance load, the nominal gain Ko is directly proportional to the value of the collector resistance Rk.

In the region of low frequencies (LF), the small capacitance Co can also be neglected, but it is necessary to take into account the resistance of the separating capacitor Cp, which increases with decreasing w. This allows us to obtain from Fig. 3.3 is an equivalent circuit of a low-frequency amplifier in the form of Fig. 3.5, from which it can be seen that the capacitor Cp and resistance Rн form a voltage divider taken from the collector of transistor VT1.

The lower the signal frequency w, the greater the capacitance Cp (1/wCp), and the smaller part of the voltage reaches the output, resulting in a decrease in gain. Thus, Cp determines the behavior of the amplifier’s frequency response in the low-frequency region and has virtually no effect on the amplifier’s frequency response in the medium and high frequencies. The greater the Cp, the less distortion of the frequency response in the low-frequency region, and when amplifying pulse signals, the less distortion of the pulse in the region of long times (decline of the flat part of the top of the pulse), as shown in Fig. 3.6.

In the high-frequency (HF) region, as well as in the midrange, the resistance of the separating capacitor Cp is negligible, while the presence of capacitance Co will determine the frequency response of the amplifier. The equivalent circuit of the amplifier in the HF region is presented in the diagram in Fig. 3.7, from which it can be seen that the capacitance Co shunts the output voltage Uout, therefore, as w increases, the gain of the cascade will decrease. An additional reason for reducing the RF gain is a decrease in the transconductance of the transistor S according to the law:

S(w) = S/(1 + jwt),

where t is the time constant of the transistor.

The shunting effect of Co will have less effect as the resistance Rк decreases. Consequently, to increase the upper limit frequency of the amplified frequency band, it is necessary to reduce the collector resistance Rк, but this inevitably leads to a proportional decrease in the nominal gain.

The block diagram of a complete low-frequency ULF amplifier is shown in Fig. 14.

Fig. 14 Block diagram of ULF.

Input stage separated from the group of pre-amplification stages, since it is subject to additional requirements for coordination with the signal source.

To reduce signal source shunting R i low input impedance amplifier R IN~ the following condition must be met: R IN~ >> R i

Most often, the input stage is an emitter follower, in which R IN~ reaches 50 kOhm or more, or field-effect transistors are used that have a very high input resistance.

In addition, the input stage must have a maximum signal-to-noise ratio, since it determines the noise properties of the entire amplifier.

Adjustments allow you to quickly set the output power level (volume, balance) and change the shape of the frequency response (timbre).

Final stages provide the required output power in the load with minimal nonlinear signal distortion and high efficiency. The requirements for the final cascades are determined by their characteristics.

1. The operation of a power amplifier for a low-impedance load of speaker systems requires optimal matching of the final stage with the total sound impedance of the speakers: ROUT~ ≈ R H .

2. The final stages consume the bulk of the energy of the power source and efficiency for them is one of the main parameters.

3. The share of nonlinear distortions introduced by the final stages is 70...90%. This is taken into account when choosing their operating modes.

Pre-terminal cascades. At high output powers of the amplifier, the purpose and requirements for the pre-final stages are similar to the final stages.

Besides this, if two-stroke the final stages are made of transistors the same structures, then the pre-terminal cascades should be phase inverted .

Requirements to preamp stages stem from their purpose - to amplify the voltage and current created by the signal source at the input to the value necessary to excite the power amplification stages.

Therefore, the most important indicators for a multistage preamplifier are: voltage and current gain, frequency response (AFC) and frequency distortion.

Basic properties of pre-amp stages:

1. The signal amplitude in the preliminary stages is usually small, so in most cases nonlinear distortions are small and can be ignored.

2. The construction of pre-amplifier stages using single-ended circuits requires the use of non-economical mode A, which has virtually no effect on the overall efficiency of the amplifier due to the low values ​​of the quiescent currents of the transistors.

3. The most widely used circuit in preliminary stages is the connection of a transistor with a common emitter, which makes it possible to obtain the greatest gain and has a sufficiently large input resistance so that the stages can be connected without matching transformers, without losing in gain.

4. Of the possible methods of stabilizing the mode in preliminary stages, emitter stabilization has become the most widespread as the most effective and simplest in circuit.

5. To improve the noise properties of the amplifier, the transistor of the first stage is chosen to be low-noise with a high value of the static current gain h 21e >100, and its direct current mode should be low-current I ok = 0.2...0.5 mA, and the transistor itself To increase the input impedance, the ULF is switched on according to a circuit with a common collector (CC).

To study the properties of preliminary amplification stages, a equivalent electrical diagram them by alternating current. To do this, the transistor is replaced by an equivalent circuit (an equivalent generator E OUT, internal resistance R OUT,pass-through capacity S K), and all elements of the external circuit that affect the gain and frequency response (frequency distortion) are connected to it.

The properties of the preliminary amplification stages are determined by the scheme of their construction: with capacitive or galvanic connections, on bipolar or field-effect transistors, differential, cascode and other special circuits.

Pre-amplification stages General information. The preamplifier amplifies the voltage or current fluctuations of the signal source to the values ​​that must be applied to the input of the final stage to obtain the specified power in the load. The preamplifier can be single- or multi-stage. Transistors in the pre-amplification stages are switched on with an OE, and the lamps are switched on with a common cathode, which allows for the highest gain. Turning on a transistor with OB is advisable when input stages, operating from a signal source with low internal resistance. To reduce nonlinear distortion in pre-amplifier stages, mode A is preferred.

• Based on the type of connection between the stages (with multi-stage amplifiers), amplifiers are distinguished with capacitive,
• transformer
• galvanic coupling (DC amplifiers).

Capacitively coupled amplifiers. Amplifiers with capacitive or CN-coupling are widely used. They are simple in design and setup, cheap, have stable characteristics, are reliable in operation, and are small in size and weight. Typical amplifier circuits using transistors and capacitively coupled tubes The frequency response of a capacitively coupled resistor stage can be divided into three frequency regions: lower low frequencies, mid midrange and upper high frequencies. In the low-frequency region, the gain Kn decreases (with decreasing frequency) mainly due to an increase in the resistance of the interstage coupling capacitor Cp1. The capacitance of this capacitor is chosen to be large enough, which will reduce the voltage drop across it. Typically, the low-frequency range is limited by the frequency fH, at which the gain is reduced to 0.7 of the mid-frequency value, i.e. Kn=0.7K0. In the mid-frequency region, which makes up the main part of the amplifier's operating range, the gain Kо is practically independent of frequency. In the high-frequency region fB, the decrease in gain Kb is due to the capacitance Co=/=Cout+Cm+Cwx (where Cwx is the capacitance of the amplifying element of the cascade; Cm is the installation capacitance, Cwx is the capacitance of the amplifying element of the next cascade). They always try to minimize this capacitance in order to limit the signal current through it and provide a high gain. Calculation of a resistor pre-amplifier stage. Initial data: amplified frequency band fn-fv = 100-4000 Hz, frequency distortion factor MH

• 1. Selecting the type of transistor. The collector current of the cascade, at which the amplitude of the input current of the next cascade is ensured Iin.tsl, Ik = (1.25h-1.5)IEx.tsl = .(1.25-7-1.5) 12= 15 -5-18 mA. Let's assume Ik = 15 mA. According to the current Ik and the cutoff frequency, which should be fashga>3fv|Zsr = 3fv(Pmin + Pmax)/2 = 3-4000(30 + 60)/2 =
• =540000 Hz=0.54 MHz, select transistor MP41 for the cascade with the following parameters: Ik=40 mA; UKe=15 V; |3min = 30; pmax = 60; famin = 1 MHz.
• 2. Determination of the resistances of resistors RK and Ra. These resistances are determined based on the voltage drop across them. Let us assume that the voltage drop across resistors R* and Re is 0.4 Ek and 0.2 Ek, respectively. We select resistors MLT-0.25 270 Ohm and MLT-0.25 130 Ohm.
• 3. The voltage between the emitter and collector of the transistor is operating point ikeo=Ek - !K(RK+Ra) = lQ - 15-10-3(270+130)=4 V. With Ukeo=4 V and Ik=15 mA according to the static output characteristics
• kam (Fig. 94, a), we determine the base current Ibo = 200 μA at the operating point O. Using the input static characteristic of the transistor (Fig. 94, b) ike = 5 V for Ibo = 200 μA, we determine the bias voltage at the operating point point O/Ubeo=0.22 V.
• 4. To determine the input resistance of the transistor at point O" we draw a tangent to the input characteristic of the transistor. The input resistance is determined by the tangent of the tangent angle
• 5. Definition of divider, bias voltage. The resistance of the divider resistor R2 is taken as R2=(5-15)Rin.e. Let's take R2=6Rin.e=6-270 =1620 Ohm. We select a resistor MLT-0.25 1.8 kOhm according to GOST. The divider current in the pre-amplification stages is taken Id = (3-10) Ibo = (3-10) -200 = 600-2000 µA. Let's assume Id = 2 mA. Resistance of resistor R1 of the divider. We select a resistor MLT-0.25 3.9 kOhm according to GOST.
• 6. Calculation of containers. The capacitance of the interstage coupling capacitor is determined based on the permissible frequency distortions Ms introduced at the lowest operating frequency. Capacitance of the capacitor Let's take an electrolytic capacitor with a capacity of 47 μF with Urab>DURE=0.2 Ek=0.2-10=2 V.

Transformer coupled amplifiers. Transformer-coupled pre-amplification stages provide better matching of amplifier stages compared to resistor-capacitively coupled stages and are used as inverses to supply a signal to a push-pull output stage. Often a transformer is used as an input device.

Circuits of amplifier stages with serial and parallel connection of a transformer are shown in. The circuit with a series-connected transformer does not contain resistor RK in the collector circuit, therefore it has a higher output resistance of the cascade, equal to the output resistance of the transistor, and is used more often. In a circuit with a parallel-connected transformer, a transition capacitor C is required. The disadvantage of this circuit is the additional loss of signal power in the resistor RK and the reduction in output resistance due to the shunting action of this resistor. The load of the transformer stage is usually the relatively low input impedance of the subsequent stage. In this case, step-down transformers with a transformation ratio n2=*RB/R"H are used for interstage communication

The frequency response of a transformer-coupled amplifier has a reduction in gain in the low and high frequencies. In the low-frequency region, the decline in the cascade gain is explained by a decrease in the inductive resistance of the transformer windings, as a result of which their shunting effect of the input and output circuits of the cascade increases and the gain K=Ko/ decreases. At medium frequencies the influence of reactive elements can be neglected. In the high-frequency region, the gain factor is affected by the capacitance of the collector junction C and the leakage inductance ls of the transformer windings. At a certain frequency, capacitance Sk and inductance Is can cause voltage resonance, as a result of which at this frequency a rise in the frequency response is possible. Sometimes this is used to correct the frequency response of an amplifier.

The amplifying mode of the transistor is determined by the constant voltages between the electrodes and the currents flowing in the electrode circuits. They are set by the elements of the external circuits of the transistor, which make up its switching circuit. The amplification device, its wiring, power source and load form amplifier stage.

Fig. 20 Diagram of an amplifier stage based on a transistor with OE

Symbols in the diagram:

R VX. V~ And R OUT V~- input and output resistance of transistor V1 to alternating current without

taking into account the elements of the external circuit (piping).

R IN.~ And R OUT~- input and output resistance of the amplifier stage.

R U- signal source resistance.

R H~- equivalent cascade load resistance to alternating current.

R VX.SL- input impedance of the next stage.

U m .ВХ- amplitude of the input signal.

U m .OUT- amplitude of the output signal.

Note: All circuit resistances are measured in the direction of the arrow when the circuit is broken along the dotted lines.

Regardless of the transistor connection circuit: with a common emitter (CE), a common base (CB) or a common collector (OC), the purpose of the elements of the amplifier stage is the same.

Let's consider the purpose of the elements of the standard wiring of a transistor connected with a common emitter (CE) in a typical amplifier stage circuit (Fig. 20).

Power supply decoupling filter R f S f.

When powering the amplifier from a rectifier, the power filter R f S F ensures smoothing of ripples of the rectified voltage of the electrical network E K .

The resistance of the resistor R Ф is selected based on the permissible reduction in efficiency. amplifier and ranges from ohm fractions in final stages up to units kOhm in low-power cascades, so that ΔU =(0,1…0,2)E K. Then the capacitance of the capacitor S F for audio frequencies can reach tens And hundredsμF, and to calculate it you can use the approximate formula

S Ф > 10⁄ (2π F Н R Ф)

Basic divider R B1 R B2.

Two resistors R B1 And R B2, connected in series according to permanent current between power bus E K and the common wire are base divisor supply voltage and form the initial base bias U 0B = U B – U E between the base and emitter of transistor V1. This is the tension U 0b determines the operating mode of the transistor: A, B or AB.

The lower the resistance of the resistors R B1 R B2 the higher the temperature stability of the cascade, but at the same time the input resistance of the cascade is unacceptably reduced variable current R IN~, for which R B1, R B2 And R VX. V~(transistor input impedance) included parallel.

R ВХ~ =(R VX. V~∙ R B) ⁄ (R VX. V~ +R B), Where R B =(R B1∙ R B2) ⁄ (R B1+ R B2)

Therefore, typical base divider resistor values ​​for preamp stages are: R B1 – tens of kOhms, R B2 – units - tens of kOhms.

Collector load resistance RK.

Resistor R K forms the flow path for the quiescent collector current I 0K, which is determined by the selected operating mode of transistor V1 (A, B or AB).

Highly resistive commutator load R K affects the amplifying properties of the transistor, since the angle of inclination of the output dynamic characteristic depends on its rating. The higher the resistance of the resistor R K(tens of kOhms) the greater the voltage gain of the cascade K U and, conversely, the less R K(hundreds of Ohms) – the greater the current gain K I.

The maximum power gain will be at comparable values R K And R OUT V~(output resistance of the transistor to alternating current).

According to AC signal, collector load resistance R K connected in parallel R OUT V~ and can lead to an unacceptable decrease in the output resistance of the cascade R OUT~ .

Auto bias resistor R E.

Transistor emitter current I E(as permanent I 0E so and variable I m E), flowing through a resistor R E forms a voltage drop across it U E. This voltage is the feedback voltage U OS, since it is related to the input parameters of the transistor by the expression: U 0B = U B – U E,

Where U B– voltage at the base of V1, measured in relation to the common wire.

As will be proven in subsequent topics, negative feedback (NF) opposes changing the parameters of the amplifier stage, ensuring stabilization of its mode, including temperature.

For example, an increase in temperature tºС causes an increase in emitter current I 0E And U E, but this automatically reduces the initial base offset U 0B = U B – U E, which turns off the transistor and, as a result, reduces the emitter current, compensating for its dependence on temperature. Hence the name R E– resistor auto offset. Thus, DC OOS has a beneficial effect on the stability of the operating mode of the amplifier stage.

But due to the flow of signal current I m E through R E OOS is formed by variable current, which, unfortunately, reduces the gain of the cascade. By connecting in parallel with the resistor R E high capacity capacitor S E, it is possible to reduce the equivalent resistance of the emitter circuit by several orders of magnitude for the lowest operating frequencies.

Capacitor S E designed to eliminate negative feedback on alternating current, as a result of which gain reduction can be avoided.

Isolating capacitors C P1 C P2– eliminate connection between cascades by permanent current In their absence, the operating modes of all transistors galvanically (directly) connected to each other will be interdependent. Moreover, a slight change in the mode of the first transistor due to the amplifying properties will lead to an unacceptable change in the mode of the last one.

The capacity of the interstage separating capacitor in ultrasonic audio frequency amplifiers reaches tens And hundreds of microfarads(µF), and the output coupling capacitor, in front of the loudspeaker - thousandsµF. In high-frequency circuits the capacitance S R decreases inversely with operating frequency. Using field effect transistor with a large input impedance, C P is sharesµF (for example 0.1 µF).

2. Operating principle of the amplifier stage(Fig.22)

In rest mode(in the absence of a signal) constant component of the collector current I 0K flows from + E K through R K, transition EC VT 1, R E, -E K. The DC component of the collector voltage, if we consider I 0E ≈ I 0K, is equal to:

U 0K = E K - I 0K (R K + R E)

In boost mode, when a signal is applied to the cascade input, the alternating component of the collector circuit current I m K flows through several parallel circuits:

1. EC VT 1 → C P2 → EB VT 2 →-E K (common wire);

2. EK VT 1 → R K → S F →-E K;

3. EK VT 1 → S р2 → R B1 → S Ф →-E K;

4. EC VT 1 → C P2 → R B2 →-E K.

Thus, the load impedance for variable signal current R n~ is the equivalent resistance parallel included R K, R B1, R B2, R VX. V 2,

R N~ =(R K∙ R IN.SL.) ⁄ (R K+R IN.SL.),

Where R VX.SL= (R VX. V 2~ ∙ R B1∙ R B2) ⁄ (R VX. V 2~ ∙ R B1 + R VX. V 2~ ∙ R B2 + R B1∙ R B2)

Fig. 22 Diagram of an amplifier stage with OE.

Only the output current component of the amplified signal is useful I m B2, flowing through the first of the listed branches, since only it will be amplified in the next amplification stage. The remaining direct and alternating currents, flowing through the transistor's binding elements, will lead to dissipation of the energy of the power source and signal, reducing the efficiency of the cascade.

The passage and processing of the signal in the circuits of the amplifier stage is clearly visible from the oscillograms at the characteristic points of the circuit shown in Fig. 22.

When a signal is applied to the input of the cascade U m .ВХ previously constant voltages in the diagram U 0B, U 0K, U 0E will become pulsating U m B, U m K, U m E, changing synchronously with the amplitude of the input signal. The oscillograms show that the signal voltages U m B, U m K, U m E, will be shifted relative to the time axis in the positive or negative region by the amount of constant potentials at these points U 0B, U 0K, U 0E, depending on the polarity of the power supply “+ E K” or “-E K”.

Only when the transistor is turned on once according to the circuit with OE, the phase of the output signal (oscillograms U m K And as a consequence U m .OUT), removed from the manifold will change by 180º. Therefore, a cascade with a transistor switched on according to a circuit with an OE is called inverse . For other switching on of the transistor with OK and OB day off And input signals always match By phase.

To determine the connection circuit of a transistor with OE, OK, OB, you must use the following rule (example for OE):

If the input signal is applied to basic transistor circuit, and the output is removed from collector, then the third electrode – emitter, is general for the input and output signal, regardless of how it is included in the circuit.

Fig. 23 and Fig. 24 show circuits with the inclusion of transistors with a common collector OK and a common base OB and their features are shown.

Fig. 23 Diagram of an amplifier stage with OK.

Important properties of an amplifier stage with a transistor connected with OK are:

1. Large entrance R BX (tens of kOhms) and small output ( tens of ohms) resistance , which improves coordination with previous and subsequent stages.

2. The input signal is not inverted, i.e. input U VX and day off U OUT the signals are in phase (φ = 0).

3. Voltage gain is less than unity ( K U< 1 , But K I >> 1).

Fig. 24 Diagram of an amplifier stage with OB.

The properties of a transistor amplifier stage with OB are opposite to the properties of a cascade with OK. Cascades with a transistor switched on according to a circuit with an OB are practically not used in low-frequency ULF amplifiers (ultrasound audio frequencies).