What are push-pull circuits of inductive converters used for? duplex converters. Computer architecture computing systems telecommunications networks

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The simplest push-pull inverter is the Royer oscillator. Here, the transistors are alternately in a state of saturation and cutoff (Fig. 5.7).

Figure 5.7 - Duple oscillator

After the power is turned on, a current flows through the resistor R 1, opening both transistors. The circuit is symmetrical and the collector currents of the transistors are equal to each other i K 1 \u003d i K 2, the self-induction EMF in the windings W 1 is also equal in magnitude, but oppositely directed. Therefore, the collector winding is generally neutral and nothing is induced in the base winding. Due to thermal, shot or flicker noise, the current of one of the transistors will instantly increase. Let i K 1 > i K 2, then an EMF will appear in the base winding, as shown in Fig. 5.7, under the influence of which VT1 opens a little, and VT2 closes slightly, i K 1 increases even more, EMF increases, etc. an avalanche-like process takes place, as a result of which VT1 enters saturation, and VT2 enters a cutoff state. The operating point of the core enters the saturation region, the current growth stops, the self-induction EMF changes sign to the opposite to support the falling current, and a reverse avalanche process occurs, as a result of which VT2 enters saturation, and VT1 enters a cutoff state, and so on.

This is a self-oscillator with a saturable transformer. The induction in the core changes from –B m to +B m . . Resistor R1 is used to start the circuit, and resistor Rb limits the base current in the open state.

Due to the finite speed of transistors operating in saturation, the collector current dissipation time is not zero and the turn-off time is greater than the turn-on time. Therefore, at the moment of changing the polarity of the voltage to W 1, VT1 does not yet have time to go into the cutoff state, and VT2 has already turned on and, to the still open VT1, voltage is applied

(5.6)

Therefore, the collector current has a surge - the so-called through current (Fig. 5.8).

Figure 5.8 - Through currents in the Royer circuit

The value of the through current can be several times higher than the operating current.

Therefore, such circuits are rarely used in modern power supplies, but in amateur radio practice it is very widespread - simplicity and reliability, with a low output power - up to 100 watts, make the circuit very attractive.

For higher powers, independent-excited converters are used to reduce power losses in the saturating output transformer. The control circuit becomes more complicated, control signals are formed with a margin of time to turn off the transistors.

Push-pull also includes bridge and half-bridge circuits. Figure 5.9a shows the power circuit of the bridge inverter, and in fig. 5.9b - diagram of work with active load. The keys work in pairs and in turn (VT 1, VT 4 and VT 2, VT 3). The losses here are greater than in the conventional circuit, since two keys are connected in series in the current circuit. The voltage on the private key is only Ek, so this scheme is preferable at high supply voltages. The voltage waveform on the load and the current waveform are the same.

Figure 5.9 - Bridge inverter

In practice, the load is rarely active, usually it has an inductive character (Fig. 5.10) and the current in the primary winding cannot change instantly.

Figure 5.10 - Bridge inverter with inductive load

After switching the keys (VT1,4 close, VT2,3 open), under the action of self-induction EMF, the current flows for some more time () through the primary winding in the same direction. Keys VT2,3 do not hold reverse voltage and can be broken through this EMF of self-induction. To protect them and create a path for the discharge current of the inductance, all switches are shunted with diodes. On fig. 5.10 conditionally shows only two of them. The energy stored in the inductance is returned to the source through the circuit: minus the source E K, diode VD3, winding W1, diode VD2, plus the source E K, regeneration takes place, and in order for the current to flow into the source, the EMF value exceeds E K by an amount. The instantaneous power on the interval is negative. (5.7)

Energy recovery can also play a positive role. For example, urban electric transport and locomotives on the railway. In them, when moving, energy is consumed from the contact network by drive motors. When braking, the motors switch to generator mode, the kinetic energy of motion is converted into electrical energy and returned to the network. In power supplies, regeneration only leads to additional losses and should be avoided. In a bridge inverter, for example, you can change the key management algorithm, as shown in Figure 5.11.

Figure 5.11 - Bridge inverter without regeneration

In this circuit, with the keys VT1 and VT4 closed, energy is transferred to the load and accumulated in the inductance. After opening VT1, the self-induction EMF changes sign, as shown in Fig. 5.11a, and the inductance is discharged through the open key VT4 and the protective diode VD3 to the load. Here, the time margin is such that the inductance is completely discharged and higher harmonics appear in the output voltage. If there is no gap between the currents i p and i 1, then there will be no dip in the output voltage and there will be less higher harmonics in its spectrum.

In inverter bridge circuits, there are four controlled keys and a rather complex control circuit. The half-bridge inverter circuit, which is shown in Figure 5.12, allows you to reduce the number of keys.

Figure 5.12 - Half-bridge inverter

Here, capacitors C 1 and C 2 create an artificial source midpoint . When VT 1 is open, C 1 is discharged to the load and C 2 is recharged, and when VT 2 is open, vice versa (C 2 is discharged to the load and C 1 is recharged). The voltage applied to the primary winding of the transformer is equal to the voltage across one capacitor.

In autonomous portable and mobile radio equipment that consumes relatively low power, low-voltage direct current sources operating independently of the external network are used as sources of electricity: galvanic cells, batteries, thermogenerators, solar and nuclear batteries. Sometimes, for the operation of radio equipment, it becomes necessary to convert the DC voltage of one rating into a DC voltage of another rating. This task is performed by various DC converters, namely: electromachine, electromechanical, electronic and semiconductor.

In a semiconductor converter, direct current energy is converted into rectangular pulse energy using a switching device. MOS FET and IGBT transistors and thyristors are used as the main elements of this device. Converters with AC output are called inverters. If the inverter output is connected to a rectifier that includes a smoothing filter, then at the output of a device called converter you can get a constant voltage U out, which can differ significantly from the input voltage U bx, , those. The converter is a kind of DC voltage transformer.

With a high value of the supply voltage, as well as in the absence of restrictions on mass and volume, it is rational to perform converters on thyristors. Semiconductor converters based on transistors and thyristors are divided into unregulated and adjustable, the latter being used as DC and AC voltage stabilizers.

According to the method of excitation of oscillations in the converter There are schemes with self-excitation and with independent excitation. Self-excited circuits are pulse self-oscillators. Circuits with independent excitation consist of a master oscillator and a power amplifier. Pulses from the output of the master oscillator are fed to the input of the power amplifier and control it.

1. Converters with self-excitation

Converters with self-excitation are performed at a power of up to several tens of watts. In radio devices, they have found application as low-power autonomous sources, power supplies, and as master generators of powerful converters. The block diagram of a self-excited converter is shown in fig. 1.

Rice. 1. Structural diagram of a voltage converter with self-excitation

A constant supply voltage is applied to the input of the converter U BX. In the oscillator, the constant voltage is converted into a voltage in the form of rectangular pulses.

Rectangular pulses with the help of a transformer change in amplitude and are fed to the input of the rectifier, after which at the output of the converter (converter) we obtain the required value and DC voltage U exit . With a rectangular pulse shape, the rectified voltage is close to constant in shape, as a result of which the smoothing filter of the rectifier is simplified.

2. Single-cycle voltage converter.

The operation of the circuit (Fig. 2), like most converters, is based on the principle of interrupting the direct current in the primary winding of a pulse transformer using a transistor operating in the key mode.

Rice. 2. Single-ended semiconductor converter

self-excited voltage

The primary winding of the transformer ω k is included in the collector circuit of the transistor, and the feedback winding ω b is included in the emitter-base circuit. Since the windings ω to and ω b are placed on the same magnetic circuit, the magnetic connection existing between them and the order of connecting the ends of the windings ultimately provide positive feedback in the oscillator.

When connecting a DC power supply U BX in the collector circuit of the transistor VT and in the winding ω to begins: a current flows, which causes an increasing magnetic flux in the magnetic circuit of the pulse transformer. This flow, acting on the feedback winding ω b, induces an EMF of self-induction in it, and the winding ω b turns on, relative to the winding ω to so that the EMF induced in it opens the transistor even more (for r-p-r transistor on the base relative to the emitter, an additional negative voltage is created). When the magnetic flux reaches saturation, the EMF and currents in the windings will disappear, a back-EMF will appear, blocking the transistor, and the process will start over. It should be noted that with an open transistor VT due to the small value of its internal resistance, the voltage drop across it will be very small, even at a current equal to the saturation current. Therefore, in this case, almost all the input voltage U BX applied to the primary collector winding of the transformer ω k.

As a result of the periodic switching on of the transistor, a current will flow through the primary winding of the transformer ω, the pulses of which will have an almost rectangular shape. Pulses of the same shape, repetition rate and polarity are transformed into the secondary winding of the transformer ω out; these pulses are used to obtain a rectified voltage using a half-wave rectifier. Resistor RR B in the base of the transistor limits the base current.

It is advisable to use converters of the described type at a high value of the output voltage U B s X and low currents, in particular, for powering a high-voltage anode in cathode-ray tubes. Main disadvantage The single-cycle oscillator circuit is the constant magnetization of the magnetic circuit, due to the fact that the current flows through the collector (primary) winding of the transformer in only one direction. ), when low efficiency is not a determining factor.

65 nanometers is the next goal of the Zelenograd Angstrem-T plant, which will cost 300-350 million euros. The enterprise has already submitted an application for a soft loan for the modernization of production technologies to Vnesheconombank (VEB), Vedomosti reported this week, citing Leonid Reiman, Chairman of the Board of Directors of the plant. Now Angstrem-T is preparing to launch a line for the production of chips with a 90nm topology. Payments on the previous VEB loan, for which it was purchased, will begin in mid-2017.

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Timing diagrams

When choosing a scheme for constructing a switching power supply, the developer is primarily guided by the expected overall dimensions and the simplicity of circuit solutions. Network sources that feed loads of low power (up to 100-150 W), built into fairly large equipment, are best built according to a single-cycle fly-back scheme. For stabilizers that do not require galvanic isolation of the load from the mains, a chopper circuit is used. When powered by galvanic cells or batteries, you can use a booster circuit. However, situations are not ruled out in which the listed converters and stabilizers cannot be used.

Case one- the device, powered from the AC mains, has limited dimensions (for example, it is not possible to place a sufficiently large storage transformer of the fly-buck converter in the instrument case).

Second case- - the power consumption of the device exceeds 150...200W.

Third case- Separate parts of the device circuit require additional power, galvanically isolated from the rest of the circuit.

In all these cases, the development of so-called two-stroke circuits of converters having galvanic isolation of primary and secondary circuits. The most widespread among push-pull converters are three schemes: two-phase push-pull (push-pull), half-bridge (half-bridge) and bridge (full-bridge). The advantage of these schemes is that, if necessary, the developer can easily introduce an output voltage stabilization unit into the design, or refuse it. In the first case, the converter will be a full-fledged power source to which any load can be connected. In the second case, a simple electrical energy converter will be obtained, requiring additional output stabilization. In some cases, such a simple converter will suit the developer. Since all three push-pull converter circuits have many analogies, we will talk about them in one chapter, focusing on individual features and conducting a comparative analysis.

Push-pull two-phase circuit

Rice. 14.1. Basic push-pull converter circuit

This circuit (Fig. 14.1) consists of two key elements, which are used as powerful bipolar or field-effect transistors. The Tr transformer has primary and secondary windings divided into half windings. The output of the power source is connected to the midpoint of the primary winding. The secondary circuit is a two-phase full-wave rectifier VD1, VD2, as well as a ripple filter (in this circuit, the filter element is the capacitor C f).

In the first cycle, as shown in Fig. 14.2, l is closed, Kl2 is open, the current flows through half-winding 1.1 and is transformed into half-winding 2.1. Diode VD1 is open and conducts current i 2.1, recharging the capacitor Cf. In the second stroke, shown in Fig. 14.3, the key Kl.l closes and the key Kl2 opens. Accordingly, the current i 1.2 flows through
half-winding 1.2 and is transformed into half-winding 2.2. The diode VD1 is locked, the diode VD2 conducts current i 2 2, recharging the capacitor C f.

Thus, the transfer of energy to the load is carried out during both cycles.

To move on to the parameters of real circuits, we first assume that we still have the possibility of using ideal elements. That is, the transistors can instantly switch, there is no reverse recovery time of the diodes, the primary winding has a very large value of the magnetization inductance (according to the equivalent circuit). Under these conditions, it is very easy to determine the dependence of the output voltage on the input value. The voltage of the primary winding is transformed into the secondary winding without loss, with a transformation ratio:

Transformation ratios nl And p 2 they are assumed to be the same, moreover, they equalize the number of turns of the primary and secondary half-windings:

The voltage on the primary winding in the closed key mode (without taking into account the voltage drop on the power key):

Since the circuit is built with full-wave rectification at the output, the relationship between supply voltage and load voltage is:

So far, it is not entirely clear to us how to introduce voltage regulation at the load. Therefore, it is necessary to remember the duty cycle and extend it to the push-pull circuit. Let's try to figure out what happens if we narrow the control pulses, as shown in Fig. 14.4. The fill factor in the case of a push-pull circuit is determined in exactly the same way as for a single-cycle circuit:

where γ is the ratio of the open state time of one key to the switching period.

Rice. 14.4. To the determination of the fill factor

In this case, we determine the duty cycle for one arm of the push-pull circuit. . Let us determine the average value of the load current, taking into account that the energy transfer is carried out during both half-cycles, which means that the average voltage value for one cycle of operation must be doubled:

Rice. 14.5. Graphs explaining the operation of the push-pull converter circuit

Thus, by adjusting γ in the range from 0 to 0.5, it is possible to linearly adjust the voltage across the load. In a real circuit, in no case should the converter be allowed to work with γ = 0.5. A typical value of γ should not exceed 0.4...0.45. The thing is that the elements used cannot have ideal properties. As we know, the primary winding has a limited inductance L μ, which accumulates energy:

The maximum current i μ, shown on the graph (Fig. 14.7), is determined from the relationship:

When Kl1 opens, the energy accumulated in the magnetic circuit tends to maintain the current. If the circuit did not have a protective diode VDp 2, shown in fig. 14.6, a negative voltage surge would occur on CL2. The ability of bipolar transistors to withstand negative voltage surges is small (a few volts), so the discharge current i μ must be closed through the VDp 2 diode. The diode practically “short-circuits” the winding ω 2 2 and quickly discharges L μ (Fig. 14.8). During the discharge, thermal energy is released, which can be taken into account through the following relation:

Rice. 14.6. To the explanation of switching

processes in a real push-pull scheme

converter Rice. 14.7. Determination of the magnetizing current

Rice. 14.8. Discharge magnetizing inductance

When the push-pull converter is operating, the discharge diodes are switched on alternately. It should also be remembered that MOSFET transistors, as well as some IGBT transistors, already have these diodes, so there is no need to introduce additional elements.

The second trouble is related to the finite recovery time of the rectifier diodes. Imagine that at the initial time the diode VD1 conducts current. The directions of action of the EMF are shown in the diagram "a" (Fig. 14.9).

Rice. 14.9. Explanation of the effect of the finite recovery time of rectifier diodes

When the transistor VT1 is turned on, the EMF changes direction (circuit "b"), the VD2 diode opens. But at the same time, the VD1 diode cannot close instantly. Therefore, the secondary winding turns out to be a shorted diode pair VD1-VD2, which causes current surges in the key element (this is clearly seen in the equivalent circuit of the transformer). The shape of the current of the primary winding on the combined graph at y \u003d 0.5 will be the same as shown in Fig. 14.10.

Rice. 14.10. The nature of the current of the transformer windings in the case of the presence of ideal and real rectifier diodes

In order to avoid switching surges, it is necessary, firstly, to introduce a pause between the closing of CL1 and the opening of Cl2 for a time not less than twice the diode reverse recovery time tgg. Secondly, if possible, it is better to abandon conventional diodes and use Schottky diodes.

The voltage on the closed key transistor is the sum of the supply voltage U n and EMF of the primary half-winding, which is currently open. Since the transformation ratio of these windings is 1 (windings with the same number of turns), the overvoltage on the key transistor reaches 2 U n . Therefore, when choosing a transistor, you should pay attention to the allowable voltage between its power electrodes. It should also be taken into account that the current of the key transistor is the sum of the direct load current, converted into the primary circuit, and the linearly increasing current of the magnetization of the primary winding inductance. The current has a trapezoidal shape.

When determining the maximum duty cycle in the case of using FETs that switch fairly quickly, one must be guided by the value of the diode reverse recovery delay. Time period during which switching is prohibited:

∆t ass= 2trr.

Fill factor correction:

Maximum fill factor:

When using bipolar transistors and IGBT transistors, the maximum possible duty cycle is reduced due to the turn-off and decay time of these transistors, as well as the characteristic "tail":

Experience shows that 1 fill factor does not exceed 0.45 in the most favorable case.

What is the difference between a real scheme and an ideal one? The resistances of the open diode and the key transistor are different from zero. It is possible to take into account the voltage drop across these elements (and correction for the transformation ratio) as shown in Fig. 14.11.

a) Rectifier diodes: in the open state, an average of 0.7 ... .1.0 V drops on the diode (standard diode), or 0.5. ..0.6 V (Schottky diode);

b) Key transistors: if a bipolar transistor or an IGBT transistor is used as a key, the voltage Uke will drop on the key (in saturation mode). A typical saturation voltage value is 0.2. ..0.5 V. For a MOSFET transistor, you need to calculate the voltage:

Preliminary calculation of the main parameters of the push-pull converter circuit should determine the transformation ratio P and overall power of the transformer. We have already found out that:

Otherwise (taking into account the voltage drop across the switches and rectifier diodes):

Where - minimum possible supply voltage (set at the beginning of development).

For example, if a battery-powered converter is being designed, this voltage could be the voltage measured across the battery terminals at the end of its life.

It is also necessary to determine the minimum value of the duty cycle γ min , based on the maximum value of the supply voltage (this parameter will be needed when determining the parameters of the smoothing output filter):

Now we can move on to determining the overall power of the transformer, which is calculated as half the sum of the power transferred to the primary winding and received from the secondary windings. In the case of a two-winding transformer, the overall power can be determined as the sum of the load powers and the power consumed by the control circuit (if the converter is built in such a way that the control circuit is powered by the same transformer):

The choice of the required magnetic circuit for the transformer is carried out according to the formula for the overall power, derived in the section "How the transformer works". Using this formula, we must determine the product SS 0 . It should be noted that for push-pull converters it is preferable to use toroidal magnetic circuits, since the transformers wound on them are the most compact. So, the overall power of a transformer wound on a magnetic core of specific dimensions:

Where η tr- Transformer efficiency (typical value 0.95...0.97) The following condition must be met by the developer:

The number of turns of the primary half-winding can be found by the following formula, which is a form of writing the law of electromagnetic induction:

Number of turns of the secondary half-winding:

After that, you need to select the required wire diameter and check the filling of the window with copper. If the coefficient a turns out to be more than 0.5, it is necessary to take a magnetic circuit with a large value of S 0 and recalculate the number of turns.

The overheating temperature of a transformer can be determined using the following formula:

where ∆ E n - - overheating (T n \u003d T a +∆T n);

T p- transformer surface temperature;

R p- total heat loss (at the active resistance of the winding and in the magnetic circuit);

S cool -- the area of ​​the outer surface of the transformer;

α - heat transfer coefficient (α \u003d 1.2 10 -3 W / cm 2 ° С).

After calculating the transformer, it is necessary to select the power elements according to the permissible values ​​​​of currents and voltages, to facilitate, if necessary, the thermal regime with the help of heat-removing radiators.

A very important issue that now needs to be considered is the choice of a control circuit for a push-pull pulsed source. Not so long ago, all these circuits had to be designed on discrete elements, which gave rise to rather cumbersome and not very reliable solutions. Micro assemblies used to control single-cycle regulators and converters are not directly suitable for use in push-pull circuits, since you need to have two paraphase outputs controlled by one generator. In addition, the microcircuit must contain a special unit for guaranteed limitation of y in order to prevent emergency situations and through currents. It is desirable to have additional safety shutdown inputs. Recently, a large number of specialized microcircuits have been developed, which already have almost all the necessary nodes.

The TL494 chip (manufactured by Texas Instruments, has a domestic analogue of KR1114EU1), which is widely used to control power supplies for computers such as IBM-PCs, is described in detail in an accessible book. As an example, consider the no less interesting CA1524 chip manufactured by Intersil. This microcircuit contains in its composition control and monitoring circuits, it functions normally when powered from 8 to 40 V. It can be used as part of any stabilizer and converter circuits described in this book.

The main components of the microcircuit (Fig. 14.12):

Thermally compensated reference voltage source 5 V;

Accurate RC oscillator;

Error amplifier (difference between the required load voltage and the real voltage at the output of the stabilizer);

Comparator of the key transistor control circuit;

Error amplifier for the current signal in the primary circuit;

push-pull output stage built on fast bipolar transistors;

Remote on/off control circuit.

Rice. 14.12. Functional units of the CA1524 chip from Intersil

Pulse-width regulation (PWR) was considered by us in the chapter on the chopper circuit of the stabilizer. In this case, the WIR scheme works in exactly the same way. The only feature is the flip-flop and the logic circuit that "routes" the control pulses, alternately directing them to one output (transistor Sa), then to the other (transistor Sb). The trigger is synchronized with clock pulses from the master oscillator. Clock pulses have a certain duration, which serves to organize a protective pause between turning off one power transistor and turning on the second one. Thus, the duty factor ymax cannot be more than 0.45 (the total pause time for the two outputs is 10%). The dead time can be adjusted by selecting the appropriate value of the time-setting capacitor St. The frequency of the master oscillator is determined by the ratio of rt and St (the choice of these elements, shown in Fig. 14.13, is carried out from the graph, Fig. 14.14). It can be seen that tangible values ​​of the pause time are obtained at sufficiently large capacitance values ​​St. If the elements of the timing circuit have already been selected, the "dead time" can be adjusted within 0.5 ... 5.0 µs by connecting the capacitor Cd to pin 3, as shown in fig. 14.15. The value of this capacitor is in the range of 100 ... 1000 pF. However, the scheme developers recommend using this method only as a last resort.

Rice. 14.13. Elements of the frequency setting circuit Rice. 14.14. Graph of the selection of elements of the timing chain

Another way to control dead time is to limit the voltage of the error amplifier (Fig. 14.16).

The error amplifier (pins 1, 2, 9) has a gain of 80 dB (10000) and can be reduced to the required value by connecting a resistor R L between pins 1(2) and 9 (depending on whether the direct or inverting switching circuit is used by the developer of the pulse source). Error amplifier unity gain frequency f-- 3 MHz. Chip designers note that an error amplifier not covered by a feedback circuit has a so-called transfer characteristic pole at 250 Hz

(the phase shift between the input and output signal at this frequency reaches 45 degrees). The pole is clearly visible on the graph (Fig. 14.18). This is another reason why an amplifier should not be used without the feedback circuits shown in Fig. 14.17.

Rice. 14.15. Additional capacitor Q, which regulates the "dead time" (a), and the schedule for choosing its value (b)

Rice. 14.16. Method for adjusting dead time by limiting the voltage value of the error amplifier

Rice. 14.17. Feedback in the error amplifier

A source without feedback can turn into a generator. To eliminate the possibility of self-excitation, it is recommended to connect a correction circuit to pin 9, as shown in. rice. 14.19.

Rice. 14.18. AFFC of the error amplifier Rice. 14.19. Corrective circuit that eliminates self-excitation

Parameters of the CA1524 chip:

Supply voltage 8...40 V;

The maximum frequency of the master oscillator - 300 kHz;

Output voltage instability - no more than 1%;

Temperature instability - no more than 2%;

Capacitance range St - 0.001 ... 0.1 μF;

Resistance range rt - 1.8 ... 120 kOhm;

Input offset of the error amplifier - 0.5 mV;

Error amplifier input current - 1 μA;

The maximum voltage "collector-emitter" transistors Sa and Sb -40V;

Current protection is triggered when the current consumption of the microcircuit exceeds 100 mA;

The rise time of the collector current of transistors Sa and Sb -0.2 μs;

The collector current decay time of transistors Sa and Sb is 0.1 µs.

The microcircuit also has an external control input (pin 10). Shutdown occurs when a high level is applied (nominal current 0.2 mA).

We will return to the CA1524 chip during the practical development of an experimental push-pull converter, and now we will consider low-power integrated sources built according to the push-pull scheme that have appeared recently. The need for a low-power converter appears when it is necessary to obtain a voltage whose source has no galvanic connection with the rest of the circuit. For example, digital devices for transmitting information over long lines need such sources. Interference induced in a long line can damage the transmitting and receiving devices, so the communication line is decoupled using matching transformers or optoelectronic devices. Active line matching devices require power.

The second example of using galvanically isolated sources is much closer to the subject of the book. A little later, we will consider the so-called bootstrap method of controlling push-pull cascades. We will see that in this circuit we need a source that is galvanically isolated from the common wire. In dynamic mode, this function, as it turns out, can be successfully performed by a capacitor. But in static mode, you can’t do without a normal source. More recently, this problem was solved with the help of an additional; windings on the network transformer, which, of course, did not contribute to reducing the dimensions of the circuit. The advent of miniaturized transducers gracefully solved this problem.

For example, let's analyze the device of the DCP0115 chip from the company] Burr-Brown, the functional units of which are shown in fig. 14.20, and the appearance - in fig. 14.21. The microcircuit contains a high-frequency generator and a push-pull cascade that works; with a frequency of 400 kHz. A miniature transformer is connected to the power stage, which, nevertheless, allows you to get 1 W of power at the load (at an output voltage of 15 V). There is also a soft start circuit and a thermal block circuit with the ability to recover after a shutdown. Synchronization pins (sync in, sync out) are used when the microcircuit works in conjunction with other pulse sources available in the device. Synchronization allows you to avoid frequency beats and reduce radiated radio interference. The microsource is made in a DIP-14 package.